The present invention relates to a switching power supply for supplying a stabilized DC voltage to industrial and consumer electronic appliances and to a control method for the same. More particularly, the present invention relates to improvements in the stability of a switching power supply comprising a plurality of switching power supply circuits.
In recent years, as electronic appliances are made more inexpensive, compact, efficient and energy saving, switching power supplies being inexpensive, compact and efficient and having output stability are demanded strongly as power supplies for use in these electronic appliances. In particular, in the case of power supplies for supplying electric power to semiconductor devices, as semiconductor devices are made more highly integrated, power supplies having higher stability at a lower voltage and capable of supplying a larger current are demanded strongly. In a switching power supply circuit in a switching power supply, an AC voltage having a rectangular waveform is generated by switching devices that repeat ON/OFF operation. The voltage is then changed to a desired AC voltage by using a high-frequency transformer and converted into a DC voltage by using a rectifier circuit and a smoothing circuit. The transformer for use in this switching power supply has a configuration wherein a primary winding and a secondary winding thereof are obtained by winding a wire on a magnetic substance by a plurality of times. Voltages applied to and induced in the windings are changed by adjusting the number of turns thereof. Generally speaking, in a switching-power supply circuit, a rough voltage change is carried out by the transformer, and a fine voltage adjustment is carried out by the PWM control of the ON/OFF ratios of the switching devices. The numbers of turns of the primary winding and the secondary winding of the transformer are determined mainly by a voltage to be applied. The higher the voltage, the more the number of turns required. The more the number of turns of each winding of the transformer, the larger the volume of a portion required for insulation between the windings. As a result, there is a problem of increasing the outer dimensions of the transformer.
A voltage nearly proportional to an input voltage is applied to switching devices in a switching power supply circuit. In the case when the input voltage is high, a high voltage is applied thereto. A semiconductor device is mainly used as the switching device. In the case of a semiconductor device in which the voltage applied at the OFF time is high, the resistance and the voltage drop at the ON time are generally large. As a result, the loss in the semiconductor device increases, and radiating means for dissipating the heat due to this loss is made larger, whereby it is difficult to make the apparatus more compact. To solve this problem, a configuration is devised wherein each input side of a plurality of switching power supply circuits are connected in series so that the voltages applied to the respective switching devices are lowered.
The series connection system on the input sides of a plurality of switching power supply circuits in a conventional switching power supply is known in Official Gazette of Unexamined Patent Publication No. Sho 62-138061.
FIG. 4 is a circuit diagram showing a configuration example of a conventional switching power supply wherein the input sides of a plurality of switching power supply circuits are connected in series. In FIG. 4, an input DC voltage from an input DC power supply 201 is supplied across input terminals 202a and 202b. The series circuit of a plurality of capacitors 203, 204, 205 and 206 is connected across the input terminals 202a and 202b. The input DC voltage applied across the input terminals 202a and 202b is divided by the respective capacitors 203, 204, 205 and 206. In the descriptions given below, the plurality of capacitors 203, 204, 205 and 206 connected across the input terminals 202a and 202b are referred to as a first capacitor 203, a second capacitor 204, a third capacitor 205 and a fourth capacitor 206, respectively. The series circuit of a first switching device 207 and a second switching device 208 is connected across both ends of the series circuit of the first capacitor 203 and the second capacitor 204. Furthermore, the series circuit of a third switching device 209 and a fourth switching device 210 is connected across both ends of the series circuit of the third capacitor 205 and the fourth capacitor 206.
A first transformer 211 has a primary winding 211a, a first secondary winding 211b and a second secondary winding 211c. One end of the primary winding 211a is connected to the connection point of the first capacitor 203 and the second capacitor 204, and the other end of the primary winding 211a is connected to the connection point of the first switching device 207 and the second switching device 208. The first secondary winding 211b and the second secondary winding 211c are connected in series.
A second transformer 212 has a primary winding 212a, a first secondary winding 212b and a second secondary winding 212c. One end of the primary winding 212a is connected to the connection point of the third capacitor 205 and the fourth capacitor 206, and the other end of the primary winding 212a is connected to the connection point of the third switching device 209 and the fourth switching device 210. The first secondary winding 212b and the second secondary winding 212c are connected in series.
An anode of a first rectifier diode 213 is connected to the first secondary winding 211b of the first transformer 211, and an anode of a second rectifier diode 214 is connected to the second secondary winding 211c. Cathodes of the first rectifier diode 213 and the second rectifier diode 214 are connected to each other. As described above, the first rectifier diode 213 and the second rectifier diode 214 are connected to the first transformer 211, thereby rectifying the voltages generated in the first secondary winding 211b and the second secondary winding 211c. 
As shown in FIG. 4, one end of the series circuit of a first choke coil 215 and a smoothing capacitor 216 is connected to the connection point of the first secondary winding 211b and the second secondary winding 211c. The other end of this series circuit is connected to the connection point (cathodes) of the first rectifier diode 213 and the second rectifier diode 214.
An anode of a third rectifier diode 217 is connected to the first secondary winding 212b of the second transformer 212, and an anode of a fourth rectifier diode 218 is connected to the second secondary winding 212c. Cathodes of the third rectifier diode 217 and the fourth rectifier diode 218 are connected to each other. As described above, the third rectifier diode 217 and the fourth rectifier diode 218 are connected to the second transformer 212, thereby rectifying the voltages generated in the first secondary winding 212b and the second secondary winding 212c. 
One end of a second choke coil 219 is connected to the connection point (cathodes) of the third rectifier diode 217 and the fourth rectifier diode 218, and the other end thereof is connected to one end of the smoothing capacitor 216. The smoothing capacitor 216 is connected across output terminals 220a and 220b, and a load 221 connected across the output terminals 220a and 220b consumes electric power.
As shown in FIG. 4, the voltage generated at the output terminal 220a on the positive side is input to one end of an error amplifier 223, and the reference voltage from a reference power supply 222 is input to the other end of the error amplifier 223. The error amplifier 223 compares the output voltage across the output terminals 220a and 220b with the reference voltage of the reference power supply 222 and amplifies the error therebetween.
A triangular wave generation circuit 224 generates a reference triangular wave signal serving as a reference for generating a PWM signal supplied to each of the first switching device 207 to the fourth switching device 210. The generated reference triangular wave signal is input to one end of a comparator 225. The comparator 225 compares the reference triangular wave signal with output of the error amplifier 223, thereby generating the PWM signal. The PWM signal generated by the comparator 225 is alternately distributed by a distributor 226 to the two output terminals thereof, thereby driving each of the first switching device 207 to the fourth switching device 210.
The operation of the conventional switching power supply configured as described above will be described referring to the operation waveform diagram of FIG. 5.
In FIG. 5, a waveform A shown in a part of (a) is a waveform of the output signal from the error amplifier 223, and a waveform B shown in the part of (a) is a waveform of the output signal from the triangular wave generation circuit 224. A part of (b) in FIG. 5 shows a waveform of the output signal of the comparator 225. A part of (c) in FIG. 5 shows a waveform of the drive signal of the first switching device 207 and the third switching device 209. A part of (d) in FIG. 5 shows the waveform of the drive signal of the second switching device 209 and the fourth switching device 210. A part of (e) in FIG. 5 shows a waveform of the applied voltage of the first switching device 207, and a part of (f) shows a waveform of the applied voltage of the second switching device 208. A part of (g) in FIG. 5 shows a waveform of the applied voltage of the primary winding 211a of the first transformer 211 and the primary winding 212a of the second transformer 212, and a part of (h) shows a waveform of the current of the first choke coil 215 and the second choke coil 219.
As shown in parts of (c) and (d) in FIG. 5, the first switching device 207 and the second switching device 208 are operated at a phase difference of 180 degrees therebetween by drive signals from the distributor 226, thereby turned ON/OFF at nearly the same duty ratio so as not to be turned ON simultaneously.
When the first switching device 207 is in the ON state, the voltage of the first capacitor 203 is applied to the primary winding 211a of the first transformer 211. When the second switching device 208 is in the ON state, the voltage of the second capacitor 204 is applied to the primary winding 211a of the first transformer 211. Furthermore, when the first switching device 207 is in the ON state, the voltage obtained by addition of the voltage of the first capacitor 203 and the voltage of the second capacitor 204 is applied to the second switching device 208 (see the part of (f) in FIG. 5). When the second switching device 208 is in the ON state, the voltage obtained by the addition of the voltage of the first capacitor 203 and the voltage of the second capacitor 204 is applied to the first switching device 207 (see the part of (e) in FIG. 5).
When both the first switching device 207 and the second switching device 208 are in the OFF state, the voltage of the first capacitor 203 and the voltage of the second capacitor 204 are applied to them, respectively.
The change in the applied voltage during the ON/OFF operation of the third switching device 209 and the fourth switching device 210 is similar to the above-mentioned change in the applied voltage during the ON/OFF operation of the first switching device 207 and the second switching device 208.
When it is assumed that duty ratios of the first switching device 207 to the fourth switching device 210 are nearly the same, the applied voltages of the first capacitor 203 to the fourth capacitor 206 become nearly the same, that is, ¼ of the input DC voltage, respectively. Hence, only the half of the input DC voltage is applied to the respective switching devices 207, 208, 209 and 210. In addition, only ¼ of the input DC voltage is applied to the primary windings 211a and 212a of the transformers 211 and 212.
Voltages generated in the secondary windings 211b and 211c of the first transformer 211 and the secondary windings 212b and 212c of the second transformer 212 are rectified by the first to fourth rectifier diodes 213, 214, 217 and 218, and smoothened by the first choke coil 215, the second choke coil 219 and the smoothing capacitor 216.
Only during the ON periods of the first to fourth switching devices 207, 208, 209 and 210, a voltage represented by (¼)·(Ns/Np)·Vin is generated in the secondary windings 211b and 211c of the first transformer 211 and the secondary windings 212b and 212c of the second transformer 212. Herein, Np designates the number of turns of the primary winding 211a of the first transformer 211 and the number of turns of the primary winding 212a of the second transformer 212. Ns designates the number of turns of the secondary windings 211b and 211c of the first transformer 211 and the number of turns of the secondary windings 212b and 212c of the of the second transformer 212. In addition, Vin designates the value of the input DC voltage. Hence, the output voltage value after smoothing can be adjusted by adjusting the ON periods of the first to fourth switching devices 207, 208, 209 and 210, and-by changing the product of the voltage applied to the first choke coil 215 and the second choke coil 219 and the time of the voltage application.
The output voltage is compared with the reference voltage of the reference power supply 222 by the error amplifier 223. The error obtained by the comparison is amplified and then compared with the reference triangular wave signal by the comparator 225 and fed back to the PWM signal. In this way, in the conventional switching power supply shown in FIG. 4, the output voltage is fed back, and the output is stabilized.
In the conventional switching power supply wherein the input side DC connection system is used as described above, the voltage applied to each switching device is a half of the input voltage, and the voltage applied to the primary winding of each transformer is ¼ of the input voltage. Therefore, in half-bridge converters, the applied voltage of each switching device and the applied voltage of the primary winding of each transformer can be halved approximately. As a result, in the conventional switching power supply, switching devices with low breakdown voltages can be used, and the numbers of turns of the windings of the transformers can be decreased.
Next, current mode control being used as a control method in a conventional switching power supply will be described.
FIG. 6 is a circuit diagram showing a case wherein the current mode control is applied to a switching power supply having a step-down converter. In FIG. 6, the input DC voltage from an input DC power supply 201 is supplied across input terminals 202a and 202b, and a capacitor 227 is connected across the input terminals 202a and 202b. The series connection of a first switching device 228 and a second switching device 229 is connected to the capacitor 227. The first switching device 228 and the second switching device 229 repeat ON/OFF operation alternately.
As shown in FIG. 6, one end of a choke coil 230 is connected to the connection point of the first switching device 228 and the second switching device 229, and the other end of the choke coil 230 is connected to a smoothing capacitor 231. The choke coil 230 and the smoothing capacitor 231 are connected in series, and the smoothing capacitor 231 is connected across output terminals 232a and 232b. Electric power is supplied to a load 233 connected across the output terminals 232a and 232b. 
In the conventional switching power supply configured as described above, when the first switching device 228 is in the ON state, the input voltage is applied to the series circuit of the choke coil 230 and the smoothing capacitor 231. When the second switching device 229 is in the ON state, the series circuit of the choke coil 230 and the smoothing capacitor 231 is short-circuited.
As shown in FIG. 6, the voltage generated in the output terminal. 232a on the positive side is input to one end of a first error amplifier 235. The reference voltage from a reference power supply 234 is input to the other end of the first error amplifier 235. The first error amplifier 235 compares the output voltage across the output terminals 232a and 232b with the reference voltage of the reference power supply 234, amplifies the error therebetween and outputs the amplified error to a second error amplifier 237. A current detector 236 detects the current flowing in the choke coil 230 and outputs the detected current to the second error amplifier 237. The second error amplifier 237 compares the output of the first error amplifier 235 with the output of the current detector 236, amplifies the error therebetween and outputs the amplified error to a comparator 239. The comparator 239 compares the reference triangular wave signal from a triangular wave generator 238 with the output of the second error amplifier 237 and generates a PWM signal. This PWM signal determines the ON period of the first switching device 228 and drives the first switching device 228. An inverter 240 inverts the PWM signal from the comparator 239 and drives the second switching device 229.
Next, an operation of the conventional switching power supply configured as shown in FIG. 6 will be described.
When the state averaging method is used, it is assumed that by the series circuit of the first switching device 228 and the second switching device 229, an amount of the input voltage Vin, corresponding to the duty ratio D thereof, is applied to the series circuit of the choke coil 230 and the smoothing capacitor 231. Hence, state equations represented by the following equations (1) to (3) are established. Herein, vout designates an output voltage, and iL designates an output current (choke coil current). Furthermore, the Laplace transforms of the output current iL and the output voltage vout are assumed to be I and V, respectively.
                                          ⅆ                          i              L                                            ⅆ            t                          =                                            1              L                        ⁢                          v              out                                +                                                    V                in                            L                        ⁢            δ                                              (        1        )                                                      ⅆ                          v              out                                            ⅆ            t                          =                                            -                              1                CR                                      ⁢                          v              out                                +                                    1              C                        ⁢                          i              L                                                          (        2        )                                          s          ⁡                      (                                                            I                                                                              V                                                      )                          =                                            (                                                                    0                                                                              -                                              1                        L                                                                                                                                                        1                      C                                                                                                  -                                              1                        CR                                                                                                        )                        ⁢                          (                                                                    I                                                                                        V                                                              )                                +                                    (                                                                                                                  v                        in                                            L                                                                                                            0                                                              )                        ⁢            δ                                              (        3        )            
wherein, I is represented by equation (4), and V is represented by equation (5).
                    I        =                                            (                              s                +                                  1                  CR                                            )                                                      s                ⁡                                  (                                      s                    +                                          1                      CR                                                        )                                            +                              1                LC                                              ·                                    v              in                        L                    ·          δ                                    (        4        )                                V        =                                            1              C                                                      s                ⁡                                  (                                      s                    +                                          1                      CR                                                        )                                            +                              1                LC                                              ·                                    V              in                        L                    ·          δ                                    (        5        )            
As shown in equation (5), a second-order lag occurs for the output voltage, and a phase lag of up to 180 degrees is generated. However, as shown in equation (4), for the choke coil current serving as an output current, a little phase lag occurs at the resonance point. However, since the numerator is first-order, a phase lag of about 90 degrees is generated. Hence, it is understood that the PWM control of the choke coil current becomes far more stable than the PWM control of the output voltage. The current mode control in the conventional switching power supply shown in FIG. 6 uses PWM to control the current of the choke coil 230. An error signal between the output voltage and the reference voltage is amplified, and the amplified signal is used as the reference signal that is used for the control. The relationship between the choke coil current and the output voltage is represented by the following equation (6).
                    V        =                                            1              C                                      s              +                              1                CR                                              ·          I                                    (        6        )            
A current control loop in the current mode control configured as described above has characteristics that its phase lag is small, its operation is stable, and its gain can be set at a large value. Since this current control loop basically forms a first-order lag system, even when its bandwidth is made larger, oscillation due to a phase lag does not occur. By virtue of the configuration using the current mode control as described above, the characteristic of the transmission from the reference signal to the choke coil current serving as an output current has almost no lag. Hence, a loop gain of the voltage control system can be made stable by using a general PI control.
As described above, in the series connection system on the input sides of the plurality of switching power supply circuits in the conventional switching power supply, switching devices with low breakdown voltages can be used, and the numbers of turns of the windings of the transformers can be decreased. However, the conventional switching power supply has a problem with respect to the stability of the output voltage. On the other hand, in the current mode control system, the output is stable, but this system has a problem wherein switching devices having breakdown voltages depending on the input voltage must be used.
However, in the field of the switching power supply, in addition to the demand for the high stability of the output voltage, the demand for the simultaneous use of the series connection system on the input sides of a plurality of switching power supply circuits and the current mode control system is increasing. When it is attempted to simultaneously use the series connection system of a plurality of switching power supply circuits and the current mode control system, the current value of each switching power supply circuit serving as a converter must be controlled so as to conform to the reference value of the current by using the error signal between the output voltage and the reference voltage. The change of the current balance among the respective converters in the case when differences are caused among the duty ratios of the respective converters will be considered herein. It is assumed that two converters, connected in series on the input sides, are A and B, respectively, and that their duty ratios are Da and Db. In addition, when the switching devices of the converters A and B are in the ON state, the currents flowing therein are determined by the currents flowing in the choke coils of the converters A and B. The currents (primary currents) flowing in the respective switching devices are assumed to be Isa and Isb. Since the two converters A and B are connected in series, the following equation (7) is established when the state is stable.Da×Isa=Db×Isb  (7)
Hence, when the duty ratio of one converter becomes relatively larger than the duty ratio of the other converter, the balance is maintained when the primary current in the one converter becomes smaller. In other words, when the duty ratio becomes larger, operation is carried out so that the primary current of the converter becomes smaller. This operation causes a contradiction wherein the duty ratio must be made larger to increase the output current, when the whole of the converter is considered. As a result, the individual current control becomes positive feedback, and no balance is obtained. Hence, the voltage balance at the time of the series connection of the plurality of converters is lost. This causes a serious problem of applying an excessive voltage to one of the converters.
Therefore, in the conventional switching power supply, in the case when it is attempted to apply the current mode control system to the series connection system on the input sides of a plurality of switching power supply circuits, a problem wherein the balance in current and voltage is not obtained among the converters. Hence, it is impossible to attain the object of carrying out the simultaneous use of the series connection system on the input sides of the switching power supply circuits and the current mode control system inside one apparatus.